ARRL/CRRL Amateur Radio 9th Computer Networking Conference, London, Ontario Cana

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ARRL 9th Computer Networking Conference 1990

Any type of BPSK demodulator can be used. I used one based on a PLL but many of the designs in the literature use the Costas loop demodulator. Position Measurement Although the receiver described in this paper does not make the position measurement some words on this are needed. The key to the position measurement is the fact that all satellites are synchronous. This is indicated in figure 4. But to the user the epochs arrive at different times due to the different path lengths.

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Again see figure 4. These differences in path lengths are called pseudo ranges. It is the receivers job to measure these time of arrival differences w. GPS time. This involves solving a system of four equations with four unknowns given the pseudo ranges and satellite positions from the measurements and satellite data. The equations are shown in figure 4. The measurement of the path delays or pseudo ranges presents a problem for the single channel receiver such as the one described here. The single channel receiver must acquire and track four satellites sequentially as opposed to a four channel receiver which can track four satellites simultaneously.

If the user knows any one of the four unknowns an equation can be eliminated. The complexity of the sequential tracking and the solution of the equations requires a computer interface and a companion program. Also needed is a high speed digital latch to measure the path delays. At this time I have not pursued the position problem.

A "chip" is the duration or length of one bit of code.

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This was for me the toughest problem to solve primarily because this is the one aspect of a GPS receiver that is completely different from conventional receivers. The method used to obtain code lock is the Tau-Dither circuit. This is a well documented method but no actual circuits! I will leave the details of my implementation of this technique for later when the receiver hardware is discussed.

There are right now many commercially made GPS receivers on the market. I wanted to build my own receiver to 1 educate myself on receiver design, this is my first 2 Build a receiver that others could understand and if not duplicate have a good starting point 3 Be able to write this article that I hope will fill the void that currently exists in the literature on GPS and spread spectrum systems in general; a complete detailed description of an entire receiver.

A benefit of this design is the fact that it is nearly all analog. Many of the commercial receivers digitize the IF and the signal disappears into a black box of digital signal processing. The analog approach lends itself more to seeing the various and different trade offs in the design especially in the code tracking loop. And although the position problem is not addressed getting the data in my opinion is the toughest nut and has the most "fruits" to be gained. The data only receiver gives excellent insights that can be applied to other spread spectrum systems.

Some representative spectrums are shown at various points. The LI signal is received using a quadrafilar circularly polarize antenna. The signal then enters a LNA. From here the signal goes through 60 feet of RG- RG could be used to the crystal down converter. The down converter converts the spectrum at If the receiver generated code is lined up with the code from the satellite then the 2nd IF will be present otherwise the receiver sees only noise. Assuming the codes are locked the 2nd IF spectrum is as shown.

All the LO's for the 2nd, 3rd and 4th IFs are generated by division of the You might be wondering where the AGC is in this design. There isn't any. The receiver runs wide open. This is not that difficult but its probably better left to a computer to decide on AGC levels. The receiver works without it so why not? They came about in sort of a evolutionary way. I had some oscillators that were close, the simulator I built made some choices for me etc.

The choice of the 28 MHz 1st IF was based primarily on existing designs for the ham band at 1. Also just the abundance of circuits in the Ham literature at 28MHz influenced me. The choice of the other IF's was based wrongly on my early attempts to make things multiples of the code clock. Not only is this not necessary you probably are asking for trouble doing this. Needless to say the design still worked and that is what counts! Detailed Description of Receiver The rest of this paper will be devoted to the details of the LI receiver. I will start at the antenna and work through to the data demodulator.

The design has good half hemisphere coverage which is exactly what is needed for a GPS receiver. The dimensions and other details of the antenna as built are given in figure 9. If you build this antenna make sure it is twisted the right way! This antenna has a gain of about 3 dB. It also is a fairly good match to 50 ohms. The Preamp Figure 6 shows a block diagram of the preamp. The first gain stage is a NEC Gasfet This FET has a gain of about 19 dB and a noise figure of 0. The input matching network is my own design.

The element that must be tuned is the inductor in the input matching network. Tuning for maximum gain will also work at the expense of a slightly higher noise figure, but still acceptable performance. The bandwidth of this filter is approximately 40 MHz. The two air variable capacitors are tuned for max gain at MHz. The small wires serve as a capacitor coupler between resonators. The insertion loss of this filter is about 2. These RF gain blocks are easy to use and relatively cheap.

There is no tuning associated with the these two stages. These two amps combine for about 30 dB of gain. Figures 7 and 8 give the construction detail of the preamp. It is constructed on double sided G board, it is one of the few places where I used printed circuit techniques. The entire circuit is mounted in a custom made aluminum box. This is acceptable but it could be better. The cable has 6 dB of loss. In reference to figure 2 the RF input from the preamp first passes through a interdigital BPF to aid in image re jection.

The detail of this filter is shown in figure The details of this amplifier are shown in figure LO generation consists of a The LO has a power level of about 7 dBm. Figure 12 shows the two spectrums of the LO. One shows the harmonic content and the other close in purity.

The doubler and the two triplers to MHz were built dead bug style on a 4 by 8 inch piece of two sided G10 board. The filters were also enclosed in small shield boxes. Figure 13 shows the circuits of the doubler and the two triplers with detail on filter construction. Layout follows the schematic using dead bug construction techniques. The last tripler to MHz is done on a separate 2 by 6 piece of G The input is fed from the tripler via RG The printed circuit pattern in negative is shown in figure Circuit layout and detail of the interdigital filter is shown in figure The tips on construction and tuning apply but the dimensions given in figure 16 must be used.

I used end covers on the filter box which are not used in the Handbook design. Figure 17B shows the xtal downconverter system. A buffered This is used for testing but could also drive the dividers to generate the 2nd and 3rd LO's. If a This approach would eliminate the need for the additional oscillator and allow the The design used for the This shows up as undesirable Doppler which eats away at the Doppler margin.

Another undesirable is that this frequency is the same as the first IF.

If any of the This is not as bad as it would first appear. The correlation is performed by phase modulating the LO drive. As mentioned above when the receiver generated code is lined up within two chips the code on the LI signal the second IF will be present, otherwise noise. The The details of the circuits for the 2nd, 3rd, 4th IF's and associated LO's is shown in figure The ECL parts are super glued with legs flattened to G10 board. Connections should be short. The Pulse stretcher is used to lengthen the pulses from the MC The rest of the 2nd IF generation circuitry is mounted in a small shield box.

To drive the DBM phase modulator and are biased negative with respect to ground. The rest of the circuit is fairly straight forward and layout follows schematic with "dead bug" methods of construction on G10 board. The MRF 's used in both the 2nd and 3rd IF's are overkill and 31 2NA's or equivalent will work fine although the bias resistors may have to be changed.

The details are shown in figure Construction is "dead bug" on G10 with the layout following the schematic. I used a Vectron oscillator that had a kHz FM range. The frequency drift should be under Hz. Its the "pull" range that is hard to come by. The oscillator used should be able to be pulled at least 40 kHz at MHz. This oscillator can be replaced by making the See above under crystal downconverter. Both functions are implemented by controlling the code clock frequency via the VCXO.

An analog switch chooses between the two. When the switch is in Track position the system is said to be "closed loop". When in the Scan position the system is "open loop? Various waveforms and spectrums are shown accompanying the block diagram of figure Some waveforms are open loop others are close loop. The open loop waveforms are marked with an asterisk. Open Loop Operation : In order to understand closed loop operation it is first necessary to understand open loop operation. In the open loop mode the code clock VCXO is held at a constant frequency offset from 1.

This frequency difference, about 10 Hz, causes the receiver generated code to "slip" by the code on the LI signal. Ignoring the effect of the Dither for now lets look at what occurs as the two codes slide by each other. As the two codes slip they will come to point where they "correlate". Correlation is when the two codes are within two chips of alignment. Before correlation the output of the BPF is just noise. The 20 kHz IF doesn't appear all at once, rather it builds in amplitude reaching a peak when 32 the two codes are in perfect alignment.

As the two codes continue to slip the 20 kHz IF amplitude decreases until we are back to our noise output from the BPF. This process gives rise to the characteristic triangular shaped correlation pulse at the output of the Full Wave Detector. Nowletslook at the effect of the Tau-Dither. It does this by determining which side of the correlation pulse the receiver generated code is on and how far it is from the peak point.

If we know which side of the correlation pulse we are on we can determine if the receiver generated code should be advanced or retarded with respect to the received code. Knowing how far off we are tells us how far we need to move the code. This information is generated by "dithering, " or switching, between two versions of the receiver generated code. One version is delayed the other is not. The frequency of the AM is the same as dither clock frequency. The amplitude of this AM increases to a maximum and then decreases to zero when the codes are in alignment.

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As the two codes continue to slip it again grows and diminishes to zero as we pass the correlation point. This "double hump" waveform can be generated by detecting the output of the dither BPF. This is shown in the block diagram but it is not used or needed in the actual circuit. At the midpoint of the double hump, at code alignment, the induced AM goes through a degree phase shift.

The AM and its degree phase shift at code alignment is caused by the triangular shape of the correlation pulse. The dither AM is now multiplied by the dither clock reference. This recovers both the phase and amplitude information simultaneously. The output of the multiplier is lowpass filtered to give the "discriminator" error output. The degree phase change causes a polarity reversal which produces the "S" shape discriminator output.

The low pass filter after the multiplier serves the same purpose as the loop filter in the more familiar phase locked loop. Much of the analysis of the code loop can be done using the tools from phase locked loops with slight modifications. Cl osed T. The code clock VCXO is now being controlled by the error voltage from the discriminator. The discriminator voltage constantly "pushes" the receiver code in the proper direction so as to keep the two codes in lock.

When the discriminator output is positive the code should be advanced, or lower the VCXO frequency. Just the opposite for negative 33 voltages. In this manner the code clock VCXO is frequency modulated to keep the codes in alignment. With the simulator the LI code and the receiver code can be displayed on a two channel scope. By triggering on either the LI or receivers code epoch that code can be made to "stand still" on the scope. The other code will move across the screen in the direction determined by the sign of the frequency difference and at the rate determined by the magnitude of the frequency difference.

If the loop is held open all the open loop waveforms can be observed as the two codes slide through correlation. In steady state the tracking jitter can be measured. This is a direct implementation of the block diagram of figure 19 with the exception of the dither detection which is omitted as stated above. LF op amps are used for there high slew rate. This circuit was taken from P. The LED is lit for a carrier level above threshold. Threshold is set by the 10 turn pot. Dither Bandpass Filter: A single op amp active filter taken from P.

This circuit has a Q of about 5. Because of the low frequency of the dither signal type op amps work fine here. The filter is tuned to about Hz. See below. The details of operation and proper connections can be found in the Exar data book. A simple RC network at the output of the multiplier serves as the loop filter. The values of R and C will greatly effect code 34 tracking performance. Other switches could be used, even relays, with proper circuit modifications. This is a one-of-eight mix that is used as a SPDT switch. The other inputs are grounded. This switch can only handle plus or minus 5 volt signals.

Because of this a gain stage is added after the switch. The switch may be damaged if inputs exceed plus or minus 5 volts. The remaining "jitter" on the code lock is affected by the CNR, the loop filter, IF bandwidth, dither clock frequency and dither delay. Using a Tau-Dither type of code tracking loop reduces the correlated IF amplitude by about 1. This loss gets larger for longer dither delays.

Longer delays give better acquisition performance but result in more jitter. The bibliography section lists a number of references on this topic. The adjusting is done via ten turn pots. The schematic, figure 20, shows the location of each pot with the exception of the Dither frequency pot. With the exception of the Threshold pot all are of the miniature, screw type adjustment variety.

The dither clock is a timer with a pot for frequency adjustment. The tuning procedure is as follows: a simulated 20 kHz carrier is AM modulated with the dither clock. This could be done with a timer and an AND gate. The output of the Hz dither BPF and the Hz dither clock are displayed on a two channel scope. Change the dither clock frequency until the two waveforms are either in phase, 0 degrees, or degrees out of phase.

This is important. It insures that the multiplication of the 35 dither AM and the dither clock is done properly. The 0 or degree alignment depends on where the dither clock is picked off; either before or after a logic inversion. Figure 22 shows the dither clock tuning setup. This brings up another point. This will invert the error voltage polarity so that advance is now a negative error voltage and retard is now a positive voltage.

Or visa versa. Therefore a bias is needed to allow for bipolar drive. This is provided by summing a bias voltage with the error voltage from the multiplier. The bias voltage is set to approximately 6 V dc. This assumes the code clock VCXO is adjusted properly. To adjust the bias a good frequency counter is needed. The adjustment is done as follows; disconnect the dither reference signal and ground pin 3 on the XR multiplier. This should force the output of the multiplier to zero volts. Ground the IF input also just to be on the safe side.

Force the track mode by reducing the threshold adjust pot until the carrier detect LED lights. This connects the VCXO control point to the output of the bias summer. Now adjust the bias pot until the frequency of the code clock is Multiplier Offset Adjust : This adjustment compensates for any d. Ground the multiplier inputs as detailed in the Code clock bias adjustment.

Now adjust the multiplier offset pot until the voltage on pin 11 of the XR multiplier is zero volts. Threshold Adjust : Threshold adjustment is provided by a ten turn precision pot with a vernier scale. Adjustment consists of turning the pot until the carrier detect LED just starts to flicker and then backing off a little so that the LED goes out. This must be done with the entire system up and consequently is a measure of the noise floor.

This sets the level at which the system will declare that a carrier is present and correlation is assumed. The vernier scale is important. The center frequency of these filters is adjusted by a 10 turn pot. The adjustment of the IF filter is done at the same time as the Doppler filters. This is a single code version of the generator shown in figure 2. Six four bit shift registers are used to implement the two 10 bit shift registers shown in figure 2. It is hardwired for satellite vehicle 9, tap points 3 and With the addition of some exclusive or gates and switching logic other codes could be generated.

Two versions of the code are generated. One delayed, one not delayed. The delayed code is the result of feeding the on time code through eight 74L04 inverter gates. These two versions are switched back and forth to produce the dither code. The dither clock is a timer followed by a divide by 2 for square wave output. The dither frequency adjust pot is set as described above. In addition to the dither code three other waveforms can be used for phase modulation of the 2nd LO. The three are code clock, on time code and CW no phase mod. These are use for testing purposes only. The is used to select one of these modulations by the setting of SI and S2, the phase modulation select switches.

Some method must be used to reset the generator to the all ones state. This is done by the code reset button. After being debouced by the two AND gates it is synchronized to the code clock by the D flip- flop The input to the shift register is held high by this synced pulse. This loads the register with all ones. Two 4 wide and gates and two 2 wide AND gates are used for this function. The output is a 1 kHz pulse train.

Figure 26 shows the schematic of the code clock generator. The VCXO is ovenized to reduce frequency drift. This oscillator will drift less than 2 Hz at VCXO pull is about Hz. This range can be increased by increasing the cap in series with the MV diode. Following the oscillator is a low pass filter used to reduce the upper harmonics. A MAR 3 is used as a buffer amp and a pick off point is provided for frequency monitoring.

A is used to do the divide by ten operation and the 1. This was fitted with a tin cover. The heating resistors were epoxyed to the cover. The completed oscillator was then put in a tight fitting Styrofoam box.

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The power, heater and RG was used for the Heater control is provided by comparing the voltage from the LM, a precision 2. When the divided voltage from the LM is below 2. When above its off. The set temperature is controlled by the 10k ten turn pot. A good frequency counter is needed to adjust this oscillator, accurate to within a few Hz at The heater must be adjusted first. Prior to applying power to the heater control adjust the Temp. Set Pot for lowest temperature, i.

Now apply power and slowly bring up temperature to about For about 3. It is set low to allow for the d. If the trimmer cannot bring it into tune try changing the oven temp a little and readjust piston trimmer. If a higher code scan rate is desired set the oscillator even lower. The limit here is the tptal pull at the control point which as mentioned above is about Hz. With this range Hz below should be feasible and would result in about the highest scan rate that could be used with this system. The heater control circuit came from the data sheet applications for the LM sensor in the Linear Data Book from National Semiconductor.

The counter is driven by a 1. The output of the counter is converted to analog by the DAC. This results in a sawtooth waveform with a 6 minute period coming out of the DAC. The output of the DAC is summed with a bias to account for zero Doppler. This is fed to the The range of the Doppler scan is determined by the voltage divider at the DAC output amp.

All three filters will have some of the signal in them since they are about Hz apart and have approximately 1 kHz bandwidths. When this condition occurs in conjunction with correlation the carrier detect circuit is tripped. This activates both code and Doppler track circuits. Track : The output of the two Doppler filters is detected and low pass 38 filtered. The comparator subtracts these two levels to determine which filter has more of the LI signal in it.

If the LI signal is more in the lower Doppler filter the comparator tells the counter to count up. If the LI signal is more in the upper Doppler filter the comparator tells the counter to count down. Doppler zero and the Doppler filters center frequency. Doppler filter adjustment : Figure 24 shows the schematic of the active filter used for the three Doppler filters. The filter is tuned via R1 a ten turn miniature pot.

A signal generator or timer is needed as an input for tuning the filters. The lower Doppler filter is tuned to approximately The upper Doppler filter is tuned to approximately The middle Doppler filter, which is used for providing the 20 kHz IF to the code loop and data demodulator, is tuned by one of two methods. The first method uses a signal generator to find the frequency at which the comparator toggles.

This is the frequency that the middle filter should be tuned to. This method will produce adequate results.

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The second method uses a either a The middle filter should first be tuned by the first method so this really is the "fine" tuning part. Assuming a simulator is available connect a dBm Set the phase modulation select switches to CW. This allows the CW to pass "unspread" through the system and no code lock is needed. Allow the system to track the CW signal. Using a voltmeter or scope tune the middle filter for maximum voltage or amplitude. This method closes the loop and should produce a more accurate result.

A bias voltage is summed with the DAC output to compensate for this error. The most straight forward is to use a frequency counter to measure the VCXO frequency and to adjust the bias pot accordingly. If there is no 39 offset in the oscillator used in the downconverter then the VCXO frequency for zero Doppler is The voltage from the DAC must set to zero to do this adjustment.

The switch on the plus input of the bias amp is used for this purpose. Assuming that the scan rates and bandwidths remain constant CNR is the determining factor. For a CNR of about 20 dB this system will have a average acquisition time of 5 minutes. This time could be decreased by increasing the scan rates. The code scan rate would have to be increased before the Doppler scan rate can be increased.

The scan rates used here are slow compared to what could be used. Data Demodulator A block diagram for the data demodulator is shown in figure The Schematic is shown in figure This design is based on a block diagram found on P. The first step in the demodulation of this data is to hard limit the 20 kHz carrier. This removes any amplitude variation. The impulses occur at every phase transition. Since there are two phase transitions per data bit, one for the rising edge and one for the falling edge, there are two impulses per data bit.

The impulses can have either positive or negative polarity and can flip arbitrarily. To make all impulses have a positive polarity a full-wave rectifier is used. This is done with a couple of LF opamps and four diodes. This design is lifted from P. After rectification a non- inverting gain of 3 is used to increase the impulse amplitude. Now a is used to clean up the impulses and bring them to TTL levels. A oneshot with a output pulse width of 8 milliseconds is used to stretch the impulse into a pulse and add some noise immunity. The data pulse train is now divided by two in order to eliminate the two pulses per bit and to "square it up".

This is done with a D-type flip-flop. The Q output is taken as the demodulated data stream. Now that we have the data signal we still need to know when to sample the data waveform. The data clock is needed. This gives the needed 50 Hz data clock. This is done using a decade counter for the divide by ten and a for the divide by 2 to get the divide by twenty. This circuit is shown in figure Adjustment can be done when tracking a satellite. When 40 the Doppler tracker settles down and is tracking the satellite monitor pin 9 of the PLL on a scope. Now adjust the VCO frequency via the 5 kohm pot until the D.

This should insure that the free running frequency of the VCO is close to 20 kHz. Another method is to use a signal generator to generate the 20 kHz IF. This is fed to the Doppler filters and the frequency varied to find the point at which the Doppler comparator toggles.

The performance of the data demodulator could be better. Signals with CNRs below about 15 dB will produce bit errors on the order of 1 in or worse. Above this the performance is quite acceptable and good results were obtained. The detected IF level that I used as a threshold was about 2 volts. The satellite passed nearly overhead at this time, this insured good signal strength. Figure 29 shows a elevation plot of satellite vehicle 9 for Nov. The afternoon occurrence was at a low elevation angle and did not provide very good data but the receiver was still able to track it.

Figure 31 shows the received data spectrum from the satellite on the 20 kHz IF taken at the afternoon occurrence. The Doppler plot shows the sawtooth scan until the satellite is "locked on" by the receiver. The detected IF slowly rises as the satellite comes up. Zero Doppler is a little off due to some frequency drift in the downconverter. As the satellite goes back down the Doppler increases to the point where lock is broken.

Given more Doppler range the receiver should have tracked for about another hour. Once lock is broke the Doppler goes back to the scan mode and the Det. IF shows just the noise level. There is a brief "lock up" after the loss of lock but it is short lived. The Data The data was read using a HP computer that had an eight bit input port. The clock was attached to bit zero and the data to bit 1.

The data line was read on either the rising or trailing edge since it was not certain at the time which would give the best results. It turned out that the trailing edge data was the good stuff. Figure 32 is a sample of the data. The data is sent in 30 bit words. Ten words make a subframe. There are five subframes for a full frame of data. The first 8 bits of each subframe are the same. This is the preamble. After the data was read into the computer a program was used to "hunt" for the preamble pattern. When five preambles are found that are bits apart a "sync" is declared.

When the "sync" is found the program then sorts the data into 30 bit words and 10 word subframes. The subframe ID's are contained in 41 bits of the second word in each subframe. The subframe ID's are printed in a column at the right margin. Both the inverted and non- inverted are printed as the program did not check for data polarity which can be arbitrary. The data polarity can be determined from the preamble. The polarity of the data shown is correct and the left hand column of subframe ID's is the correct one. Of course one could also tell by the sequence, i. The right hand column of numbers is a count of the data bits as they where read into the computer.

The important item here is the preamble. When you see that pattern repeat every bits you know you have the data! This is as far as I have taken the data analysis. To go further requires a lot more programming and possibly computer control of the receiver. This key document is available from the GPS program office. A letter must be sent to the address given below stating name and purpose.

Unfortunately they do not have to send you a copy! I could get no firm commitment on the qualifications needed to get the document, although I was told that if my address was in Moscow I would not receive a copy. If the large corporations have access to system information shouldn't the experimenter and the entrepreneur? In particular the ARRL publications. They had examples of real circuits that worked. I would also like to thank Todd Carstenson who wrote the data capture program and provided a source of encouragement. There are many people who added bits and pieces to this endeavor, librarians, co-workers, people in industry, etc, who I would also like to say thanks.

Jessop, G6JP. Spliker, Jr. Journal of the Institute of Navigation, Vol. II, P. Spilker, Jr. AES-2, No. AES, No. For those not familiar with spread spectrum systems read this book first. Ziemer and Roger L. Peterson, Macmillan Pub. Cooper and Clare D. Coherent Spread Spectrum Systems, J. Holmes, Wiley, Sams and Co. Rohde and T. Bucher, McGraw-Hill Inc. Phase Locked Loops, Roland E. Best, McGraw-Hill Inc. Jung, Howard W.

Blood Jr. OF iT L. I74 Test? Use of both directional antennas and the UHF and microwave bands is essential to obtain efficient use of hardware and spectrum resources. The use of shorter point-to-point links and small clusters of local users can achieve dramatic increases in user information throughput. Coordination and cooperation at all levels will be necessary to make a high speed amateur network a reality.

Amateur radio avoided dying in its infancy largely due to the establishment of a low speed digital network. The American Radio Relay League was founded to further this. Now we amateurs fmd ourselves entering the information age. I believe that if amateur radio is to continue in this age it must offer relevancy and that to do this amateurs must develop and implement a significant high speed information network; the alternative is for the hobby to wither and eventually die.

Although in the past radio amateurs have led the way into new technologies and operations, in recent times we have increasingly tended to adapt our operation and pursuits to existing technologies. Amateur packet radio, which has experienced very rapid growth in the last few years, is an example of this. Our interest in packet was stimulated by seeing similar communications within the industry and military complexes.

The name given to the current link layer protocol, AX. Not only at lower layers do we see this borrowing of technology The idea of a worldwide amateur BBS system and the greater dream for a digital amateur network have followed rather than led similar existing information services in the military and commercial sectors. Certainly it is to be expected that more reuse of existing tools and methods will be required as our society and world get more complex.

It is also true that insightful adaptation of methods and technologies often results in tremendous benefits. However, as we amateurs adapt our ways to meet the changing face of technology we need to examine the peculiarities of our applications, along with our strengths and weaknesses in order to achieve the most successful results.

I believe that if we are to succeed in developing and implementing a high speed amateur network that we must examine the fundamentals of communicating information by radio as well as our own resources and strengths and then design our network accordingly. At the lowest ISO layers, physical and link, we have sought to implement packet communications bv adapting existing hardware and protocols. Telephone modem hardware went into and is still inside most TNCs. Similarly, our link layer operations are tailored after the fashion of an IEEE model.

Both of these were originally intended for wire lines, a very different environment from amateur radio. I believe that many of the problems which amateur packet is now experiencing are traceable to this mismatch of solutions and environments. A high speed amateur network requires a blend of two parts; high speed communication of information and wide area general access. High speed offers the. By 73 representing and transmitting a large amount of information digitally, the possibility for a wide range of applications exists. A digital data stream can be used to represent voice, TV, FAX, as well as computer programs, files and data.

Digital representation allows general transmission, storage and retrieval of this same data and also allows error detection and correction techniques to be used. Along with this, a wide area network can allow amateurs to communicate and share resources in new ways. Such communication and shared resources could offer relevancy in the information age and rekindle the fundamental excitement and spirit with which the hobby began.

The extent of possible applications of such a network applied to the diverse interests and pursuits of amateurs is truly staggering. High data speed is necessary for both. Even a network providing low user speed requires high speed data if a large number of users ark involved and all communication can not be carried out directly between end users. Any intermediate interconnection facilities become providers of a shared resource. A fundamental limitation to communication is noise. If this limitation were not present there would be no need for any particular transmit power, antenna gain or receiver bandwidth between two stations seeking to QSO.

Whatever transmit power was recovered by the receiving antenna could simply be amplified as required to allow detection recovery of transmitted information to take place. In actuality, signal power must be sufficient to allow separation of data from the noise power. Deep space links can effectivelv maintain lower system noise temperatures at the earth end but these are at present out of consideration for the bulk of radio amateur networking use and even if used, one end of such a link is earthbound and represents a high temperature noise source to the other.

Higher speed communication requires proportionately more signal power than lower speed. Using power to allow an increased bandwidth for a given — is more effective since channel capacity then increase linearly instead of logarithmically. This ultimately requires greater bandwidth as data rate is increased. Since total information transfer is equal to an average transfer rate over a time interval, the product of rate and time, it can be seen that the total amount of information communicated is ultimately dependent upon the amount of energy transferred, the product of average power and time, between the transmitter and receiver.

For the normal case of uniform mean noise, this energy must be enough greater than the noise energy in the same interval to allow successful data recovery. The problem of efficient use of resources in designing and implementing an amateur high speed digital network then involves finding and utilizing the most efficient techniques for conveying transmitting this information- carrying energy.

In an environment of limited resources of funding and spectrum, amateurs must use available resources optimumly if we are to build and operate a high speed network. Signal Propagation by Radio as a Function of Carrier Frequency and Distance As it is usually presented, the so-called pathloss equation shows what portion of the transmitted power gets to a distant location which is separated from it by a distance in free space.

The assumption is that the distance is great enough that both transmitting and receiving antennas are in the far field regions of the other. An isotropic antenna is an antenna which radiates uniformly in all directions. It is not physically realizable but is convenient for the sake of analysis. The portion of transmitted power recovered at the receiving end, L in is Given this representation, the amount of signal received decreases as either frequency or distance is increased, A receiving antenna serves to intercept and recover a portion of the transmitted power.

If the entire surface of a sphere of radius D were surrounded by perfect receiving antennas and if the outputs of all these antennas were totaled the sum would be the total transmitted power. No power is actually lost along a free space path. To start to make this model more like a real-world situation we can substitute for the fictitious isotropic antenna a directional antenna which can actually be constructed. Except for excessive dissipative or matching loss, a directional antenna is by definition one which has gain.

It gives gain because it causes power which the isotropic antenna would have spread evenly in all directions to be focused or concentrated in one or a feu directions while reducing it in other directions. It essentially redirects power from some undesired directions to another desired direction. Antennas of higher gain have more of this focusing ability. From the vantage point of the receiving antenna on the sphere it makes no difference whether the source is a W transmitter and an isotropic antenna or a 10W transmitter feeding a directional antenna with a gain of The field strength at a receiving antenna located in the far field would be the same in either case.

Antenna effective aperture is a measure of the useful area of an antenna. Figure 1 shows the apertures of some familiar antennas. Relative Apertures Of Some Common Antennas For simple single element antennas like a monopole or dipole, the aperture can be approximated by the area of a rectangle which is a half wavelength long and a quarter wavelength wide. It is not affected by the physical size of the conductor used to make the dipole. Adding more elements, or electrical size, makes an antenna more complex and increases the aperture relative to a dipole or isotropic antenna.

Antennas with more elements or electrical size have a relatively larger aperture on receive at the same time they have a more directional beam. Antenna Apertures and Patterns Vs. Frequency Here again are some common antennas and apertures shown to scale for two amateur bands.

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As the antenna electrical size or number of elements is increased apertures increase too but for a given electrical structure the higher frequency antennas start out with a smaller physical aperture because their wavelength is smaller. As antennas much more complex than dipoles are considered, the aperture size is increasingly related to electrical antenna complexity. However, at frequencies where the dish antenna is electrically large, where it is at least 10 wavelengths in diameter, the physical aperture is relatively constant.

Notice the dish antenna near the bottom of Figure 2. It has the same physical size on each band, although its electrical size, measured in wavelengths, is about 3 times as large on MHz as it is on MHz. It has about the same physical aperture on both bands although its gain, directivity and its ability to focus a transmitted signal is about 10 times greater at the higher frequency.

There is no longer any frequency dependence to the equation, only a distance dependence. This arrangement of using a constant physical antenna size instead of a constant electrical antenna type makes a lot of sense in practice. Almost always the limitations to amateur antennas at the ham shack or at a high level site are in terms of antenna physical size rather than antenna electrical size. Antenna and tower wind loading, rotor capability and antenna size are constraints much more often than number of elements or antenna dimension measured in wavelengths.

In exactly the same way that the receive antenna size is more likely to be physically rather than electrically constrained, so is the transmit antenna size. In fact we are likely to want to use the same antenna for both receiving and transmitting since most of our communications will need to be two-way.

In this third rendition of the equation we transmit as well as receive using an antenna of constant physical size. The transmitter power is better focused to go only toward the receive antenna and not elsewhere, as frequency is increased. Once again there is a frequency dependence in the equation but this time instead of things getting worse as frequency is increased, as was the case with constant electrical antenna size, with constant physical antenna size the amount of transmitted signal reaching the receiver increases with increasing frequency.

In fact, an increase in distance incurs no additional reduction in recovered power if frequency is increased by the same amount. In summary? First Online: 16 March This process is experimental and the keywords may be updated as the learning algorithm improves. IEEE Kleinrock L, Tobagi FA: Packet switching in radio channels: part I—carrier sense multiple-access modes and their throughput-delay characteristics.

Google Scholar. CrossRef Google Scholar. Tinnirello I, Choi S: Temporal fairness provisioning in multi-rate contention-based Golestani SJ: A self-clocked fair queueing scheme for broadband applications. Wang and G. Personalised recommendations.

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